Device for transmitting or emitting high-frequency waves

ABSTRACT

The present invention relates to a device for transmitting or emitting high-frequency waves, which includes: a microstrip line ( 10 ) with one end ( 10 ′) in a substrate ( 11 ) for transmitting high-frequency useful signals; a first ground surface ( 12 ) and a second ground surface ( 13 ), which are provided on opposite sides of the microstrip line ( 10 ), for forming a TEM waveguide assembly; an opening ( 14 ) in the first ground surface ( 12 ) located at a predefined distance (d) from the end of the microstrip line ( 10 ′) for decoupling a high-frequency signal; a feedthrough device ( 15 ) for conductively connecting the first ground surface ( 12 ) with the second ground surface ( 13 ) on the lateral periphery of the microstrip line ( 10 ); and a planar coupling device ( 16 ) for receiving and transmitting or emitting the high-frequency useful signal. The feedthrough device ( 15 ) is configured in such a way that at a given frequency (f) it prevents the propagation of waveguide modes and excitation of waveguide mode resonance in the useful frequency band (F).

BACKGROUND INFORMATION

The present invention relates to a device for transmitting or emittinghigh-frequency waves.

Devices for emitting electromagnetic waves, such as planar antennaelements, which are excited using a slot aperture for producingoscillation and, therefore, emitting high-frequency waves, have becomewidespread in radio link technology, satellite communicationstechnology, and radar technology. They are used preferably in themicrowave range, since this allows small component sizes and, therefore,simple realizations at low cost.

A common planar antenna device is presented with reference to FIG. 6A,in which a slot coupling is excited via a microstrip line (MSL) 10. Tothis end, microstrip line 10 has an abrupt end 10′ and therefore formsan open-ended line. A slot 14 is located in a ground surface 12separated by a substrate 11, perpendicular to microstrip line 10, at adistance d of approximately 1/4 of the line wavelength from abrupt end10′ of microstrip line 10. Passage, i.e., coupling, of the magneticfield, which is at a maximum at this point, takes place through saidslot. This field, which is also provided with an electrical fieldcomponent, excites a planar antenna element 16—also called a patchelement—to produce sympathetic vibration and nearly complete emission ofhigh-frequency energy with a main direction of propagation which isorthogonal to ground surface 12. FIG. 6B shows a top view of thecross-section of the device according to FIG. 6A.

The disadvantage of this arrangement is that microstrip line substrates11 become very thin at higher frequencies, e.g., 254 μm in a short rangeradar application (SRR) at 24 GHz, and do not have adequate structuralstability to be expanded upon. For this reason, these substrates 11 mustbe joined with a rigid carrier material 18, as shown in FIG. 7A. Forreasons of cost, this carrier material 18 is not suitable for use inhigh-frequency applications. Carrier material 18 is placed above groundsurface 12 with a permanent connection therewith, and a cost-intensiverecess 19 must be created in carrier material 18 to ensure that theantenna is capable of functioning in the region of coupling slot 14 orantenna element 16, so that antenna element 16 can beelectromagnetically coupled via coupling slot 14.

To feed single antenna 16, a further conventional embodiment of aslot-coupled antenna uses a “buried” signal-carrying line 10 with anabrupt line end 10′ which is configured in the form of a “triplate line”and excites individual antenna 16 to produce emissions, also via a slot14. Signal line 10 is located substantially plane-parallel between twoground surfaces 12, 13, whereby in the case shown in FIGS. 8A and 8B,microstrip line 10 is located closer to one of the two ground surfaces12, 13, which results in an antenna arrangement with asymmetricaltriplate feeding. In contrast, there are also arrangements withsymmetrical feeding, i.e., embedded signal line 10 is equidistantbetween outer ground surfaces 12, 13. The symmetrical or asymmetricaltriplate arrangement has the advantage that larger line elements can behidden in a lower layer as buried structures, to reduce component size.When larger antennas are to be realized in particular which are composedof a large number of such individual antennas 16 in order to increasethe directivity of the antenna, locating high-frequency linearrangements in layers located further downward make compact assembliespossible, since the feeding network of an antenna array takes up asignificant portion of the required installation space.

Moreover, a buried feeding network does not negatively influence theemission characteristics of an arrangement of this type, in contrast to“open” distribution and feeding networks, in particular, which make aconsiderable contribution to parasitic emissions. Another advantage isthe possibility of providing easily manufactured, multilayerarrangements, since their single layers have good high-frequencyproperties and carry the particular line structures to be buried. Whensuitable layer or substrate materials are used, such as ceramics, theconnection with an additional mechanical carrier can be eliminated,since the multilayer arrangement has adequate structural stability. Lowtemperature co-fired ceramic (LTCC) substrates are particularlywell-suited for use in this field.

The antenna arrangement described with reference to FIGS. 8A and 8B hasthe disadvantage, however, that the release of waves from an abrupt end10′ of signal-carrying, center line 10 of the triplate structure isgreatly enhanced. A considerable portion of the signal power can thendisadvantageously propagate in substrate material 11, e.g., in the formof parallel plate modes or waveguide modes. If the multilayerarrangement is mounted laterally in a metallic carrier or housing, theexcitation of waveguide modes is further enhanced. The propagation ofwaveguide modes is determined by their limiting frequency fg, the valueof which depends directly on the distances from the bordering metallicwalls.

The following relationship applies in general: Limiting frequency f_(g)of a waveguide mode is shifted toward lower frequencies when thedistance from electrically conductive, e.g., metallic, walls isincreased. At the same time, the number of modes capable of propagatingin a certain frequency band increases continually. If modes of this typeare now excited in substrate 11 by open-ended line ends, the poweremitted via antenna element 16 is reduced and couplings with othercircuit parts within substrate 11, e.g., further antenna elements, areenhanced. This has a disadvantageous effect on the antennacharacteristics and the overall system behavior.

ADVANTAGES OF THE INVENTION

Compared to the known means of attaining the goal of the invention, thedevice according to the present invention for transmitting or emittinghigh-frequency waves having the features of claim 1 has the advantagethat excitation of substrate or waveguide modes in a slot-coupledantenna arrangement with a symmetrical or asymmetrical triplate line isprevented or reduced to an extent which is no longer relevant to thebehavior of the antenna or the system, without negatively influencingthe basic mode of operation of a slot-coupled emission device.

The device according to the present invention enables a cost-effectiveimprovement of the functionality of the antenna, since suppression ofthe described excitation of substrate or waveguide modes contributes toan improvement in the antenna efficiency and, therefore, systembehavior.

The idea on which the present invention is based essentially consists ofproviding a shielding measure in the region of the signal line and thecoupling slot, and of adjusting its dimensions to fulfill therequirements of each.

In other words, a device for transmitting or emitting high-frequencywaves is provided which includes a microstrip line provided with an endin a substrate for transmitting high-frequency useful signals, a firstground surface and a second ground surface which are provided ondiametrically opposed sides of the microstrip line to shield themicrostrip line, an opening in the first ground surface at a predefineddistance from the end of the strip line for decoupling a high-frequencysignal, a feedthrough device for conductively connecting the firstground surface with the second ground surface on the periphery of themicrostrip line to shield the same (e.g., using “via holes”), and aplanar coupling device for receiving and transmitting the high-frequencyuseful signal, whereby the feedthrough device is structured and/ordimensioned such that, at a given frequency of the useful signal, nowaveguide modes occur in the substrate which are capable of propagationor resonance.

Advantageous further developments and improvements of the deviceindicated in claim 1 are provided in the subclaims.

According to a preferred further development, the structure of thefeedthrough device widens in the region of the coupling opening. Thisresults in the advantage that coupling to an antenna element (patch) isnot hindered by the shielding feedthrough device in the region of thecoupling opening.

According to another preferred further development, a distance a betweendiametrically opposed feedthrough devices in the region of themicrostrip line is less than the quotient c₀÷(2·f·{square root}{squareroot over (ε _(r) )}) whereby C ₀ stands for the speed of light in avacuum, ε_(r) stands for the dielectric permittivity of the substrate,and f stands for the frequency of a useful signal. This advantageouslyprevents the following from being formed: A first waveguide mode—whichis capable of propagating—of a rectangular waveguide, which isapproximately what is present here (TE₁₀ mode), a mode with a transverseelectrical (TE) field, as viewed in the cross section.

According to another preferred further development, the followingrelationship exists between the width B between diametrically opposedfeedthrough devices in the region of the coupling opening and length Lof the feedthrough device in the region of the coupling opening:$L < \frac{1}{{\sqrt{\left( \frac{2 \cdot f_{res} \cdot \sqrt{ɛ_{r}}}{c_{0}} \right)}}^{2} - \left( \frac{1}{B} \right)^{2}}$whereby C₀ stands for the speed of light in a vacuum, ε_(r) stands forthe dielectric permittivity of the substrate, and f_(res) stands for aresonant frequency of an excitable waveguide mode which is to beprovided above a useful signal frequency band. This is an advantage interms of defining the dimensions for the feedthrough or via walls in theregion of the coupling slot, since it prevents undesired resonancefrequencies from forming cavity resonance effects within the shieldingwalls in the region of the coupling slot.

According to another preferred further development, the resonantfrequency is approximately a few percent higher than the useful signalfrequency band. Resonance phenomena are reliably prevented in thismanner.

According to another preferred further development, the device foruseful signals is designed with a frequency band between 20 GHz and 30GHz. The device is suitable for use in a SRR (short range radar)application, for example.

According to another preferred further development, the feedthroughdevice is composed of discrete feedthrough elements which are locatedlaterally adjacent to each other, preferably forming a wall which actsas an electromagnetic shield. This provides the advantage of goodshielding with feedthrough elements which are economical to manufacture,whereby the distance is selected depending on the frequency.

According to another preferred further development, the discretefeedthrough elements are round and/or cylindrical in shape. A simplemanufacturing procedure can be ensured as a result.

According to another preferred further development, the feedthroughdevice is a continuous wall. This provides the advantage of a closedshielding device, e.g., in the form of a metallic layer, which preventsvirtually all electromagnetic in-coupling and decoupling.

According to another preferred further development, the feedthroughdevice is made continuous in the region longitudinally adjacent to theend of the strip line. Advantageously, the strip line is completelyshielded.

According to another preferred further development, the feedthroughdevice is provided with a gap in the region longitudinally adjacent tothe end of the strip line. As a result, virtually no electromagneticradiation is emitted or absorbed, and manufacturing outlay is reducedslightly.

According to another preferred further development, the microstrip lineis located closer to the ground surface with the coupling opening thanto the other ground surface in the substrate, or vice versa. Thisprovides the advantage of an asymmetrical structure, which is necessary,e.g., when coupling a further microstrip line via the coupling opening.

According to another preferred further development, the microstrip lineis located nearly equidistantly between the ground surface with thecoupling opening and the other ground surface in the substrate. Thisprovides the advantage of a simple arrangement.

According to another preferred further development, the planar couplingdevice forms a second microstrip line in another plane which isprovided—with galvanic separation—to electromagnetically couple-in thisadditional microstrip line. In this manner, a signal transmissiondevice, with galvanic separation, is advantageously provided.

According to another preferred further development, the two microstriplines are substantially identical in configuration, and they overlap inthe longitudinal direction by a two-fold predefined distance, whichpreferably corresponds to nearly half the wavelength of the couplinguseful signal. Maximum electromagnetic coupling between the twomicrostrip lines is therefore ensured.

According to another preferred further development, the coupling openingis arranged parallel to the ground surface in the shape of a slot and/orrectangle. This makes it possible to develop a simple layout of thecoupling opening in the ground surface which is economical tomanufacture, and offers good decoupling and in-coupling through theslot.

DRAWING

Exemplary embodiments of the present invention are presented in thedrawing and are described in greater detail below in the subsequentdescription.

FIG. 1 shows a diagonal view of a section for explanation of a firstembodiment of the present invention;

FIG. 2 shows a diagonal view for explanation of the first embodiment ofthe present invention;

FIG. 3 shows a top view of a schematic radiation device for explanationof a second embodiment of the present invention;

FIG. 4 shows a simulation diagram for explanation of the mode ofoperation of the radiation device explained with reference to FIG. 3;

FIGS. 5A, B show a schematic illustration of a galvanically separatedcoupling device for explanation of a third exemplary embodiment of thepresent invention, whereby FIG. 5A is a longitudinal sectional view, andFIG. 5B is a cross-section along cutting plane A;

FIGS. 6A, B show a schematic illustration of a common slot-coupledplanar antenna, whereby FIG. 6A is a longitudinal sectional view, andFIG. 6B is a top view;

FIGS. 7A, B show a schematic illustration of the arrangement presentedwith reference to FIGS. 6A, B with an additional mechanicalreinforcement, whereby FIG. 7A is a longitudinal sectional view and FIG.7B is a top view; and

FIGS. 8A, B show a schematic illustration of a common slot-coupledplanar antenna with an asymmetrical triplate line feeding, whereby FIG.8A is a longitudinal sectional view and FIG. 8B is a cross-section alongcutting plane A.

DETAILED DESCRIPTION OF THE EMBODIMENTS

In the figures, the same reference numerals are used to label componentswhich are identical or have the same functionality.

FIG. 1 shows a schematic diagonal view of a slot-coupled antenna devicefor explanation of a first embodiment of the present invention.

In FIG. 1, a microstrip line 10 is embedded in a substrate 11. Thissubstrate is preferably suitable for high-frequency use and has a lowtemperature co-fired ceramic (LTCC), for example, which has gooddielectric properties with low attenuation. A first ground surface 12 isprovided above microstrip line 10, preferably parallel therewith, and isseparated by substrate 11.

The lower section of the arrangement shown is formed by a second groundsurface 13 which, identical to the first ground surface, is composed ofan electrically conductive material, preferably including a metal. Firstground surface 12 includes a coupling opening 14 which preferably hasthe shape of a rectangle and/or a slot, and which has a predefineddistance d (not shown) relative to an abrupt end 10′ of microstrip line10. This coupling opening 14 is oriented in the Y direction in thecenter of strip line 10 or the end of strip line 10′, and extends at aright angle thereto, in the shape of a cross. The predefined distance inthe X direction between slot opening 14 and end 10′ of strip line 10corresponds to nearly one-fourth of the line wavelength, i.e., λ/4, ofthe useful signal f transmitted on strip line 10, which has a bandwidthof frequency band F in the range between 20 GHz and 30 GHz in thisexample.

A feedthrough device 15 is provided between the top ground surface 12,in which coupling slot 14 is provided, and lower ground surface 13, thefeedthrough device being composed according to the present invention outof individual feedthrough elements 15′. Individual feedthrough elements15′ are preferably configured round and/or cylindrical in shape, andprovide a shielding device similar to a palisade wall.

In this case, a planar coupling device 16 serves as planar antenna,which is excited to produce resonance by the electromagnetic fielddecoupled through coupling opening 14. Planar coupling device 16 isoriented preferably parallel to coupling opening 14. The side edges ofplanar element 16, which is rectangular in this case, are also orientedpreferably parallel to the edges of coupling opening 14, i.e., in the Xand Y direction. According to the present embodiment, microstrip line 10includes an impedance transformer 17 in the region of coupling slot 14and before abrupt end 10′ of the strip line, the impedance transformerbeing used as necessary. Feedthrough device 15 widens in the region ofcoupling slot 14 and then, longitudinally adjacent to end section 10′ ofstrip line 10, comes back together again and therefore forms a closedshielding device.

A feedthrough device 15 or continuously closed shielding walls aroundstrip line 10 are suited for shielding triplate lines of this type and,consequently, preventing the formation of waveguide modes in substrate11 which are capable of propagation or resonance. Instead of providingmassive walls, it is advantageous in practice to provide feedthroughdevice 15 in the form of individual feedthroughs 15′ (via holes), which,on the high frequency side, form a nearly continuous, electricallyconductive wall by way of a sufficiently small lateral distance of thevia holes relative to each other. The maximum shielding effect isdetermined by the correct dimensioning of the distance and diameter ofthe individual feedthrough elements 15′. To now prevent waveguide modescapable of propagation or resonance, the distance separating the walls,i.e., the distance between the feedthrough device lying on one side ofstrip line 10 and the distance of feedthrough device 15 lying in the Ydirection on the other side of the strip line must not exceed a certainvalue.

The first waveguide mode—which is capable of propagating—of arectangular waveguide, which is approximately what is present here, isthe TE₁₀ mode, a mode with a transverse electrical (TE) field as viewedin the cross section. The limiting frequency of this mode is$\begin{matrix}{{f_{g} = \frac{c_{0}}{2\quad\alpha\sqrt{ɛ_{r}}}},} & (1)\end{matrix}$whereby C₀ is the speed of light in a vacuum (C₀=3·10⁸ m/s), a is thedistance of feedthrough devices 15 or via walls, and ε_(r) is thedielectric permittivity of the substrate material. It follows that theinequality $\begin{matrix}{a < \frac{C_{0}}{2 \cdot f_{g} \cdot \sqrt{ɛ_{r}}}} & (2)\end{matrix}$must be fulfilled, so that a waveguide mode is not excited up tofrequency f_(g). Distance a can be reduced depending on the electricaleffect of the shape of the via holes or their distances, and theadditional (relatively slight) influence of signal line 10.

If this via wall 15 would now be designed to follow signal line 10 inparallel at a corresponding distance a, wall 15 would intersectorthogonally oriented coupling opening 14 in the region of couplingopening 14; as a result, the mode of operation of coupling slot 14 and,therefore, the antenna or transmission device would no longer beensured. It is therefore necessary to markedly increase the distance ofvia walls in the vicinity of coupling slot 14 and then to reduce itafter slot 14 in the region of open-ended signal line 10′. The via walls15 could then be brought together after the open-ended end 10′ ofmicrostrip line 10, although this is not necessarily required, sinceexcitation of the substrate or waveguide modes would not be possible dueto the small distance present there. To achieve a maximum shieldingeffect, and to prevent electromagnetic in-couplings into the arrangementfrom the outside, feedthrough devices 15, i.e., the walls, arepreferably brought back together longitudinally adjacent to open-endedsignal line 10′.

With regard for the dimensioning or structuring of feedthrough device 15or via walls in the region of coupling opening 14, it must be taken intoconsideration that, when distance a of these walls is increased,limiting frequency f_(g) of waveguide mode becomes lower and, in fact,generally below the useful frequency f of the antenna itself. As aresult, interference of the functionality of coupling opening 14 by viawalls 15 is minimal or negligible in a draft version of the arrangement.On the other hand, this presents the risk that cavity resonance effectscan form within these shielding walls 15 with the greatly increaseddistance B in the region of coupling opening 14, the cavity resonanceeffects greatly impairing the functionality of the antenna if theseundesired resonance frequencies, which may occur, are in the usefulfrequency range. To deliberately prevent this, an appropriate length Lof via walls 15 in the X direction is selected in the region of thecoupling opening with the larger distance B of shielding walls 15 in theY direction.

In a completely closed, dielectrically filled, rectangular waveguideresonator having width B, height H and length L with ideally conductiveelectrical walls, possible discrete resonance frequencies resultaccording to the following relationship: $\begin{matrix}{f_{res} = {\frac{c_{0}}{2\sqrt{ɛ_{r}}}\sqrt{\left( \frac{p}{L} \right)^{2} + \left( \frac{m}{B} \right)^{2} + \left( \frac{n}{H} \right)^{2}}}} & (3)\end{matrix}$whereby p, m and n are whole-numbered indices, C₀ is the speed of lightin a vacuum, and ε_(r) is the dielectric permittivity of thenon-conductive filler material. For the TE₁₀ mode, which is relevanthere, m=1 and n=0. As a result, the possible resonance frequenciesdepend on width B but not on height H. The whole-numbered index p mustbe greater than zero in TE modes. This results in the first excitablecavity resonance of the TE₁₀ mode according to $\begin{matrix}{f_{res} = {\frac{c_{0}}{2\sqrt{ɛ_{r}}}\sqrt{\left( \frac{1}{L} \right)^{2} + \left( \frac{1}{B} \right)^{2}}}} & (4)\end{matrix}$In designing the antenna with slot coupling and via-hole shielding 15 ofsignal line 10, it must be noted that, although the limiting frequencyof the waveguide-like resonance according to equation (1)—whereby “a”must then be made equal to “B”—can be below the useful signal frequencyband F, but the first resonant frequency according to equation (4) mustbe above the useful signal frequency band F to prevent interference withthe mode of operation of transmission device 16 and/or the antenna.

Moreover, with the present embodiment according to FIG. 1, it should benoted that, when designing the dimensions of the shielding device orfeedthrough device 15, the use of discrete feedthrough elements 15′ witha certain lateral distance relative to each other instead of closedmetallic walls influences the limiting frequency of the waveguide modes.It must also be taken into consideration that the resonator does nothave any fully-closed walls in the region of the coupling slot, as theydo in the theoretical model, but rather large-area in-couplings anddecouplings, e.g., in the region where via walls 15 widen, whichinfluence the resonant frequency accordingly. The coupling slot 14itself also influences the resonant frequency, and open-ended signalline 10, 10′ below coupling opening 14 can change the resonantfrequency.

FIG. 2 shows a schematic diagonal view for explanation of the firstembodiment of the present invention.

A section of the arrangement according to FIG. 1 is shown in FIG. 2.Microstrip line 10 is embedded in a dielectric substrate between a firstground surface 12 and a second ground surface 13. The two groundsurfaces 12, 13 are connected with each other via electricallyconductive feedthrough elements 15′ which form a feedthrough device 15or a shielding device. According to the embodiment shown, strip line 10is arranged plane-parallel and symmetrical between the two parallelground surfaces 12, 13, i.e., in a symmetrical triplate arrangement.Strip line 10 preferably has a nearly rectangular cross section, whilethe individual laterally adjacent feedthrough elements 15 are configuredin the shape of a cylinder in particular.

FIG. 3 shows a schematic top view of an antenna device for explanationof a second embodiment of the present invention.

An antenna device according to the invention is shown in FIG. 3, wherebyit differs substantially from the embodiment shown with reference toFIG. 1 in that, in this case, feedthrough device 15 does not consist ofindividual feedthrough elements 15′, but rather of continuouselectrically conductive walls located between the first and secondground surface, providing electrical contact between the two. The usefulfrequency band F is preferably in the range of 22 GHz to 26 GHz.

The triplate structure shown in FIG. 3 is asymmetrical, i.e., thedistance from substrate 11 over signal line 10 to first ground surface12 is 150 μm, and the distance of substrate 11 below signal line 10 tosecond ground surface 13 is, e.g., 450 μm (neither of the groundsurfaces are shown in the top view according to FIG. 3). The length ofthe coupling slot, i.e., its extension in the Y direction, is 2.6 mm,for example, and the dielectric constant ε_(r) of the ceramic substratematerial is ε_(r)=7.7. For the limiting frequency of waveguide mode TE₁₀in the region of signal line 10 with small distance a from feedthroughdevice 15 or the via walls to now be above the useful frequency band F,the distance a according to equation (2) must be less than 2.46 mm, andis designed to be a=1.9 mm, for example.

To ensure that the electromagnetic coupling through coupling opening 14is not interfered with by shielding device 15, the distance of via wallsB is increased to 3.6 mm, for example, in the region of coupling slot14. As a result, the limiting frequency f_(g) of the TE₁₀ mode isreduced to approximately 15 GHz, according to equation (1). To ensurethat the first resonant frequency f_(res) of this mode is above 27 GHz,for example, which is necessary to ensure a 1 GHz-frequency distancefrom the useful frequency band F, a length L less than 2.4 mm must beselected, according to equation (4). To also compensate for theinfluences of resonant frequency f_(res) mentioned above, L ispreferably selected to be 1.2 mm in the present exemplary embodiment.

FIG. 4 shows the amplitude trace of the reflection factor as asimulation result of a full wave analysis of the entire antenna assemblyaccording to FIG. 3. Resonance clearly appears at approximately 27.7GHz, since the reflection factor has a high amplitude factor in thiscase, which corresponds exactly to the described cavity resonance effectof the TE₁₀ mode, which is followed by an analysis of associated fielddistribution images (not shown). At the same time, good reflectionattenuation occurs in the useful frequency band F between 22 GHz and 26GHz, the reflection attenuation being greater than 12 dB; above this,the matching follows a very smooth course. Based on this, interferenceby other resonance-like effects in this frequency range can be ruledout. The course of the reflection factor can be adjusted as desired inlarge regions by designing the dimensions or structures of planarcoupling device 16 or planar antenna, coupling opening 14 or couplingslot, signal line 10 and impedance transformer 17 accordingly.

FIG. 5A shows a coupling device of an electromagnetic signal withgalvanic separation. According to this third embodiment of the presentinvention, two microstrip lines 10 in a dielectric substrate 11 areseparated by a ground surface 12 with a coupling opening 14. In theillustration, lower strip line 10 extends toward the left, and has itsopen-ended end 10′ in the region adjacent to coupling opening 14, whileupper strip line 10 extends toward the right in the drawing and has itsopen-ended left end 10′ in the region adjacent to coupling slot 14. Thearrangement is configured point-symmetric to the center of coupling slot14.

The arrangement in the lower region corresponds substantially to anasymmetrical triplate feeding, which does not transmit its decoupledfield to a planar antenna (16, not shown here), however, but rather to acontinuing strip line 10. In this manner, an antenna element is notprovided, but rather a coupling device, which transmits the signal viaan electromagnetic coupling-in of a signal of a strip line in a plane toa second strip line 10 in another plane, with galvanic separation. Thefeedthrough device or shielding walls not shown in FIG. 5A have thestructures and dimensions in the region of the strip line and, inparticular, in the region of the coupling opening 14, as describedabove.

The coupling device according to FIG. 5A is shown in cross section inFIG. 5B, whereby the feedthrough device is not shown here, either, toenhance transparency, but it is still located as described above.

Although the present invention was described above with reference topreferred exemplary embodiments, it is not limited to them. Instead, itis capable of being modified in highly diverse manners.

In particular, the materials mentioned for the dielectric substrate, theground surfaces and strip line are to be regarded as examples. Moreover,the configuration of the coupling slots, the planar coupling device andthe strip line are not necessarily rectangular. Instead, they can alsohave round, oval or polygonal cross sections or top views. Thefeedthrough device and shielding walls in particular do not have toextend at a right angle to each other; instead, they can haverounded-off transitions.

1. A device for transmitting or emitting high-frequency waves with: amicrostrip line (10) provided with one end (10′) in a substrate (11) fortransmitting high-frequency useful signals; a first ground surface (12)and a second ground surface (13), which are provided on opposite sidesof the microstrip line (10), for forming a TEM waveguide assembly; anopening (14) in the first ground surface (12) located at a predefineddistance (d) from the end of the microstrip line (10′) for decoupling ahigh-frequency signal; a feedthrough device (15) for conductivelyconnecting the first ground surface (12) with the second ground surface(13) on the lateral periphery of the microstrip line (10); and a planarcoupling device (16) for receiving and transmitting or emitting thehigh-frequency useful signal; whereby the feedthrough device (15) isconfigured in such a way that at a given frequency (f) it prevents thepropagation of waveguide modes and the excitation of waveguide moderesonance in the useful frequency band (F).
 2. The device as recited inclaim 1, wherein the shape of the feedthrough device (15) widens in theregion of the coupling opening (14).
 3. The device as recited in claim1, wherein a distance (a) between diametrically opposed feedthroughdevices (15) in the region of the microstrip line (10) is less than$\frac{c_{0}}{2 \cdot f \cdot \sqrt{ɛ_{r}}}$ whereby C₀ stands for thespeed of light in a vacuum, ε_(r) stands for the dielectric permittivityof the substrate (11), and f stands for the frequency (f) of a usefulsignal.
 4. The device as recited in claim 1, wherein the followingrelationship exists between the width (B) between diametrically opposedfeedthrough devices (15) in the region of the coupling opening (14) andthe length (L) of the feedthrough device in the region of the couplingopening (14):${L < \frac{1}{{\sqrt{\left( \frac{2 \cdot f_{res} \cdot \sqrt{ɛ_{r}}}{c_{0}} \right)}}^{2} - \left( \frac{1}{B} \right)^{2}}},$whereby C₀ stands for the speed of light in a vacuum, ε_(r) stands forthe dielectric permittivity of the substrate (11), and f_(res) standsfor a resonant frequency (f_(res)) of an excitable waveguide mode whichis provided above a useful signal frequency band (F).
 5. The device asrecited in, claim 4, wherein the resonant frequency (f_(res)) has adistance greater than approximately a few percent above the usefulsignal frequency band (F).
 6. The device as recited in claim 1, whereinthe device for useful signals is designed with a frequency band (F)between 20 GHz and 30 GHz.
 7. The device as recited in claim 1, whereinthe feedthrough device (15) is composed of discrete feedthrough elements(15′) which are located laterally adjacent to each other, preferablyforming a wall.
 8. The device as recited in claim 7, wherein thediscrete feedthrough elements (15′) are round and/or cylindrical inshape.
 9. The device as recited in claim 1, wherein the feedthroughdevice (15) forms a continuous wall.
 10. The device as recited in claim1, wherein the feedthrough device (15) is made continuous in the regionlongitudinally adjacent to the end (10′) of the strip line (10).
 11. Thedevice as recited in claim 1, wherein the feedthrough device (15) isprovided with a gap in the region longitudinally adjacent to the end(10′) of the strip line (10).
 12. The device as recited in claim 1,wherein the microstrip line (10) is located closer to the ground surface(12) with the coupling opening (14) than to the other ground surface(13) in the substrate (11), or vice versa.
 13. The device as recited inclaim 1, wherein the microstrip line (10) is located nearlyequidistantly between the ground surface (12) with the coupling opening(14) and the other ground surface (13) in the substrate (11).
 14. Thedevice as recited in claim 1, wherein the microstrip line (10) includesan integrated impedance transformer (17) in the region of the couplingopening (14).
 15. The device as recited in claim 1, wherein the planarcoupling device (16) forms a second microstrip line (10) in anotherplane which is provided, with galvanic separation, toelectromagnetically couple-in this additional microstrip line (10). 16.The device as recited in claim 14, wherein the planar coupling device(16) is capable of being brought into resonance with the couplingopening (14) and is therefore capable of being excited to produceemissions.
 17. The device as recited in claim 14, wherein the couplingdevice (16) itself is capable of being brought into resonance and istherefore capable of being excited to produce emissions.
 18. The deviceas recited in claim 15, wherein the two microstrip lines are configuredsubstantially identically and overlap in the longitudinal direction by atwo-fold predefined distance (d), which preferably corresponds to nearlyhalf the wavelength of the coupling useful signal.
 19. The device asrecited in claim 1, wherein the coupling opening (14) is arrangedparallel to the ground surface (12, 13) in the shape of a slot and/orrectangle.
 20. The device as recited in claim 1, wherein the substrate () has a ceramic material, preferably low temperature co-fired ceramic(LTCC).